I. INTRODUCTION
A boost converter operating in the continuous conduction mode (CCM) is the most popular topology for implementing power factor correction (PFC) in high power applications [1]-[5]. However, the conventional operation of the CCM boost converter causes high reverse recovery losses of the output diodes as well as electromagnetic interference (EMI) noises. The switching losses and the EMI noises of the CCM boost rectifier are mainly generated during the switching transient of the main power switches. The reverse recovery current of the boost diode during the turn-on process induces a high surge current which flows through the main switch. In addition, a rapid increment of the drain-source voltage during the turn-off switching transient also results in EMI noise and turn-off switching losses. In order to increase the rated power of the CCM boost converter with a high power conversion efficiency, these phenomena should be eliminated.
Many previous studies have improved the performance of high power CCM boost converters. These studies have focused on reducing the effects of the reverse-recovery characteristic of the boost converter and increasing the power conversion efficiency. An alternative method to minimize these drawbacks is the use of a silicon carbide (SiC) diode [6]-[9] or a soft switching circuit instead of the conventional diode rectifiers. The SiC diode can reduce the power losses caused by the reverse recovery current. It has the simplest structure when compared with other methods. However, this method still has a hard switching problem of the main switch. In addition, the SiC diode has a higher forward voltage drop than that of a conventional fast recovery diode (FRD). If the output power increases, the power loss of the SiC diode is also increased. Furthermore, a SiC diode with a high rated current is more expensive than other devices. For this reason, boost rectifiers using SiC diodes are not suitable for high power applications.
The soft switching circuit for a CCM boost rectifier can reduce the reverse recovery by reducing the falling rate of the output diode's turn-off current. Various soft switching techniques using active and passive snubber circuits have been proposed [10]-[21]. The active snubber circuits can achieve zero voltage switching (ZVS) and zero current switching (ZCS) for the main switch. However, they require an additional control circuit to drive the auxiliary switch which operates under hard switching conditions. Moreover, in order to increase the power conversion efficiency for high power applications, expensive components with high power ratings are required. Therefore, the cost of the circuit is increased and the reliability is deceased by using active snubber circuits.
Generally, passive lossless snubbers are as effective as active snubbers without any additional auxiliary switches or control circuits [14]-[21]. Low cost, high performance, and high reliability are the advantages of passive lossless snubbers. A simple passive snubber circuit using a flying-capacitor has been proposed for a basic PFC topology [14], [15], a three-level PFC topology [16], and a bridgeless PFC topology [17]. These methods have basic circuit structures using snubber capacitors, inductors, and diodes. The general form of turn-on passive snubber circuits using energy recovery capability is an inductor series connected to the main switch. Therefore, these circuits require additional inductor components. In addition, these circuits cannot operate as a turn-off snubber. The turn-off snubber of the main switch is composed of resistor capacitor diode (RCD) snubbers so as to confine the rise of the switch voltage [18]. However, the power loss of the snubber resistor reduces the overall power conversion efficiency. A passive snubber circuit using saturable inductors is used to reduce the reverse recovery current of the output diodes [19], [20]. However, this causes extra voltage stress on the main switches and increases costs due to the additional inductors. A snubber circuit with saturable inductors is adapted to reduce the reverse recovery current of the output diodes [21]. It does not introduce extra voltage or current stress on the main switch. However, it needs nine additional passive components with three additional snubber inductors, which has the demerits of a large circuit size and extra cost.
In this paper, a novel passive lossless snubber circuit is proposed to improve the power conversion efficiency of a CCM boost PFC rectifier for high power applications. In order to use the proposed turn-on snubber circuit, the turn-on current of the main switch resulting from the reverse recovery current of the output diode is limited. In addition, the turn-off lossless snubber can reduce the voltage spike of the main switch during turn-off transitions. The proposed snubber circuit consists of nine additional components composed of six components for a turn-on snubber, and three components for a turn-off snubber. The proposed turn-on snubber consists of coupled inductors merged into the boost inductor. However, the additional cores of the snubber inductor are not required because three inductors are coupled in a single core. Actually, seven additional components are used for the proposed lossless snubber circuit. Therefore, it can reduce both the circuit size and the extra cost. Even though it requires a relatively large number of passive components when compared with the SiC diode-based PFC, the proposed technique in high power rectifier applications is a good solution for reducing reverse recovery problems. The performance of the proposed lossless snubber circuit will be experimentally verified using a 3.3 kW prototype PFC rectifier.
II. PROPOSED PASSIVE LOSSLESS SNUBBER
A. Circuit Diagram
Fig. 1 shows a schematic of a PFC boost rectifier with the proposed lossless snubber circuit. The conventional boost PFC converter contains a boost inductor L1, a main switch S, an output diode Df, and an output capacitor Co. The proposed turn-on passive snubber consists of two coupled inductors L2 and L3, three auxiliary diodes Dx1, Dx2, and Dx3, and an additional capacitor Cx. The inductors L1, L2, and L3 are coupled using a single core structure. The turn ratio of the coupled inductors is n:1:1 (n >> 1). The turn-off snubber circuit consists of a capacitor in parallel with the main switch Cs and two diodes Ds1 and Ds2. A PFC circuit using the proposed snubber method has been presented in international conference [22]. However, this paper did not suggest an entire power stage design. In addition, the effect of the proposed coupled inductor was not considered. In this paper, the performance improvement and analysis of the operating effects of the coupled inductors will be presented. The detailed circuit operation of the proposed PFC rectifier will be discussed in the next section.
Fig. 1.Proposed PFC boost converter with lossless snubber.
B. Operating Principles
To analyze the steady state operation of the proposed PFC converter, following assumptions are made during one switching period:
There are eight operating modes in a single switching cycle. The circuit operations in the positive half period of the input voltage are shown in Fig. 2. The dark lines denote the conducting paths for each state. The theoretical waveforms of the proposed rectifier are given in Fig 3.
Fig. 2.Operating mode of the proposed rectifier: (a) Mode 1, (b) Mode 2, (c) Mode 3, (d) Mode 4, (e) Mode 5, (f) Mode 6, (g) Mode 7, (h) Mode 8.
Fig. 3.Theoretical waveforms of the converter.
Mode 1 [t0 – t1]: At t0, S turns on. When S turns on, Df is not immediately turned off because of its reverse recovery process. Snubber inductors L2 and L3 limit the rising rate of the switch current. The inductor currents are shown as follows:
where the voltage across the switch current is Is = IL3 = IL1 - IL2, and IL1 (t0) - IL2(t0) = 0. The equivalent coupled inductance Leq will be defined in Section III.
At t1, IL3 reaches Iin. Thus, the duration of Mode 1 can be calculated in (4).
Mode 2 [t1 – t2]: The reverse recovery phenomenon of Df is finished at t1. Since Df is turned off, lL2 charges Cx through L3, Cs, and Ds2. Each inductor current and the voltage of the snubber capacitor in Mode 2 can be derived as follows:
where This mode ends at t2 when VCs (t2) = 0.
Since the turn ratio of the coupled inductors L2 and L3 is 1:1, the voltage across the snubber capacitor at t2 is VCx (t2) = VL2 (t2) - VL3 (t2) = 2VL2 (t2). The voltage across the snubber inductor L2 at t2 can be calculated in (9).
The capacitance of Cx should be much larger than the capacitance of Cs for the soft switching of the main switch S. Thus, using (8) and (9), the duration of Mode 2 can be approximated in (10).
Mode 3 [t2 – t3]: At t2, Ds2 is turned off, and Dx1 is turned on. During this period, the energy stored in L2 and L3 is transferred to Cx through the resonant path L2 - L3 - Dx1 - Cx. The boost inductor current IL1, the resonant current IL2, and the voltage across Cx are shown as follows:
where Since the coupling between inductors L1 and L3 has a turns ratio of n:1, the voltage of the boost inductor, VL1, is
At t3, IL2 is decreased to zero. Therefore, by using (12), the duration of Mode 3 is can be derived in (14).
Mode 4 [t3 – t4]: At t3, switch S is still turned on. During this period, L3 becomes short-circuit. Thus, the switch current Is equals IL1. The duration of Mode 4 defines the duty cycle of the main switch. This mode ends when S is turned off softly at t4.
Mode 5 [t4 – t5]: At t4, IL3 flows through Ds1 and charges Cs. The main switch S is softly turned off. The resonant current IL3 and the snubber voltage VCs are shown as follows:
where
At the same time, the energy stored in Cx begins to discharge through Dx2. IL2 and VCx are shown as follows:
where
At t5, the energy stored in Cs is fully charged and the voltage across Cs is VCs ≃ Vout. Therefore, by using (16) the duration of Mode 5 can be derived in (19).
Mode 6 [t5 – t6]: At t5, Cx is still discharged through Dx2 and VCx has the same value as (18). The stored energy in L3 is transferred to the output capacitor through Dx3. IL3 can be expressed as:
where ir4 is the same as IL3(t5) in Mode 5. VCx is discharged to zero at t6.
Mode 7 [t6 – t7]: At t6, the output diode Df is turned on. The energy in the input voltage source and boost inductor L1 starts to flow to the output through Df. During this period, the energy remained in L3 is fully discharged. IL3 is the same as (20). The duration of this mode can be calculated in (21).
Mode 8 [t7 – t8]: At t7, Df is still turned on and snubber diode Ds3 is turned off. During this period, snubber inductor L2 becomes short-circuit. This mode ends when the main switch S is turned on again and the operation proceeds to Mode 1.
III. ANALYSIS AND DESIGN GUIDES OF THE PROPOSED LOSSLESS SNUBBER CIRCUIT
A. The Coupled Inductor
As shown in Fig. 1, two snubber inductors of the proposed lossless snubber are built on the same core with a boost inductor. The inductance of the two snubber inductors are the same (L2 = L3), since they have the same turns ratio. However, the two inductors can no longer be considered as two individual inductors because of the coupling effect. The performance of the proposed lossless snubber circuit is strongly related to the direction of the windings. There are two similar equivalent circuits for each of the coupling formats during Mode 1, as shown in Fig. 4. The two snubber inductors can be directly coupled or inversely coupled due to the different direction selections between the two windings. In order to analyze the operation of the proposed snubber circuit through the difference in the direction of the windings, it is assumed that the coupling effect between the two inductors is perfectly coupled and that the input voltage of the snubber circuit is fixed by a constant voltage source Vin.
Fig. 4.Equivalent circuits of two coupled inductors in Mode 1. (a) inverse coupling, (b) direct coupling.
Fig. 4(a) shows the recommended coupling methods. It has inverse coupled windings. When S turns on during Mode 1, Df is not immediately turned off because of its reverse recovery characteristics. A momentary equivalent circuit of the proposed snubber circuit can express the simple circuit as shown in Fig. 4. During this period, the voltage across L2 and L3 is shown as follow:
where the coupling inductance and k is coupling coefficient. Since the inductance of L1 is much greater than the inductance of L2, IL2 should be much larger than IL1. For this assumption, the switch current Is is approximately equal to IL1. (22) can be substituted by (23).
where the equivalent inductance Leq using the inverse coupled winding can be defined by (24).
Since the coupled inductors L2 and L3 have the same turn ratio, the voltage of the inductors is also the same. At this time, the snubber inductors L2 and L3 limit the rising rate of the switch current. Therefore, by using (23), the current of snubber inductors IL2 is can be derived in (25).
If the coupling coefficient in the inverse coupled windings case is defined as k = 1, the equivalent inductance Leq is four times larger than L2.
In the case of Fig. 4(b), the equivalent inductance Leq using the directly coupled winding can be defined by (26).
If the coupling coefficient in the directly coupled windings is defined as k=1, the equivalent inductance Leq is zero. These windings cannot limit the rising rate of the reverse recovery current. Therefore, the coupling method of Fig. 4(a) is suitable for the proposed lossless snubber circuit.
B. The Passive Resonant Component
The inductance of the coupled L2 and L3 is determined by considering the variation of the output diode current in Mode 1. The current of the output diode equals the current of L2 at t0. Thus, the inductances of L2 and L3 are shown as:
where In order to reduce the current stress caused by the reverse recovery of the diode rectifier, the variation of dIL2(t0)/dt should be lower than [13]. The snubber inductors L2 and L3 should be selected with respect to (28).
The snubber capacitor Cx is continuously charged by the energy from Cs through Ds2 in Mode 2. Since the energy in Cs is completely transferred to Cx in this period, the following condition can be obtained as shown in (29).
where the voltage across Cs is VCs ≃ Vout and the ratio between Cx and Cs is α = Cx/Cs. The minimum voltage of VCx(t2) is determined by the chosen value of α. The energy stored in the coupled inductors of L2 and L3 is transferred to Cx in Mode 3. The capacitance of Cx can be designed as follows:
where Irr is the peak value of the reverse recovery current. The minimum capacitance of Cx can be obtained from (30), and this design criteria guarantees the soft switching operation of power switch S, which reduces the switching losses and improves power conversion efficiency of the proposed PFC circuit.
IV. POWER LOSS ANALYSIS
In order to improve the power conversion efficiency, reducing the effects of the reverse-recovery characteristic is one of the most important research topics in high power CCM boost PFC applications. A fast recovery diode (FRD) has a small reverse recovery time (trr). In the operation of the CCM boost PFC, trr causes the reverse recovery current to increase additional power losses and EMC problems. These drawbacks can be minimized by employing SiC diodes or soft switching circuits instead of conventional diode rectifiers. In theory, the SiC diode has a very small trr. In addition, the SiC diode which has the simplest structure can reduce the power losses caused by the reverse recovery current. However, it still has a hard switching problem of the main switch. Furthermore, the SiC diode has a higher forward voltage drop than that of the other conventional FRDs. If the output power increases, the power loss of the SiC diode is also increased.
In order to reduce these problems, a soft-switching technique with the couple inductor method has been proposed in this paper. Even though, it requires a relatively large number of passive or active components, the proposed technique in high power rectifier applications is a good solution for reducing the reverse recovery problems. In order to verify the usefulness of the proposed circuit, a detailed power loss analysis of the proposed CCM boost PFC will be discussed in the next subsections.
A. Consideration of the Conduction Losses
The major conduction losses in PFC rectifiers are caused by the three parts of the PFC's power stage; full-bridge rectifier diodes, a main PFC switch, and a boost rectifier diode. In order to analyze the conduction loss of the PFC rectifier, its circuit operation assumes that the boost inductor L1 is sufficiently large to operate under the CCM and that the AC input current is perfectly sinusoidal. In this case, there is no difference in the conduction losses of the full-bridge rectifier diodes between the lossless snubber circuit and the SiC diode. However, the calculations of the conduction losses in the main PFC switch and the boost rectifier diode should consider the effects of the lossless snubber circuit and the SiC diodes.
Generally, the conduction loss in the main MOSFET can be calculated by using the rms current passing through its drain-source channel resistance, RDS.on. In addition, the conduction loss in the boost rectifier diode can be calculated by using the forward voltage drop of the boost diode, VF. The rms current of the MOSFET can be calculated using the following equation:
where the average turn-on period of the main PFC switch, Dav, can be determined by as follows:
For this calculation, the peak current, Iin.P and the average current, Iin.av of the AC input can be derived as follows:
where η is the power conversion efficiency. Using (31) - (34), the conduction losses in the main PFC switch and boost rectifier diode can be calculated as follows:
where the conduction loss is Pcon, and VF is the forward voltage drop of the boost rectifier diode. A MOSFET (IXKR47N60C5) is used for the main PFC switch with a low Rds.on of 45mΩ, a FRD DSEI2X101-06 is used for the boost rectifier diode with a low VF, and a SiC diode (C3D20060D) manufactured by Cree is used for replacing the lossless snubber circuit and FRD. Since Rds.on is the same in the case of the lossless snubber circuit and the SiC diode, Pcon is expected to be only affected by VF. Fig. 5 shows the conduction losses of the PFC rectifier with the proposed lossless snubber and the SiC diode. Under the 10 kW load condition, the difference in the conduction losses between the lossless snubber circuit and the SiC diode in the PFC rectifier is 15 W. Therefore, the PFC rectifier with the proposed lossless snubber circuit can increase the power conversion efficiency when compared with the case of the SiC diode in high power applications.
Fig. 5.Conduction loss of a main PFC switch and the boost rectifier diode according to the proposed lossless snubber and the SiC diode.
B. Consideration of the Switching Losses
The operating waveforms of the PFC rectifier for a single switching cycle are shown in Fig. 6. When the main PFC switch turns on, in the case of the FRD rectifier, the diode cannot immediately turn off because of its reverse recovery process during trr. This simultaneous high reverse recovery current causes an additional switching loss in the main PFC switch. In order to analyze this phenomenon, the reverse recovery characteristics of the FRD are illustrated in Fig. 6 (a). IRR is the maximum reverse current of the FRD. ta is the time duration between the zero crossing point and the peak point of the reverse current, which is related to the charge stored in the depletion region of the junction. tb is time duration of the charge stored in the bulk semiconductor material. trr can be expressed as trr = ta + tb. In addition, the reverse recovery charge QRR and the maximum reverse current IRR can be calculated as shown in (36) and (37).
Fig. 6.The operating waveforms of the PFC rectifier for a single switching cycle. (a) FRD diode-based PFC, (b) The proposed lossless snubber PFC.
From (37), IRR is directly proportional to QRR. The values of QRR and trr are provided on the data sheet of the diode used in the rectifier.
A large IRR increases the current stress in the low side switches and decreases the EMI performance of the rectifier. If a SiC diode is used for the boost rectifier diode, trr in the SiC diodes is theoretically zero and IRR should also be zero. Even though these values are not completely zero in practical operation, IRR is significantly reduced when comparing to the FRD operation. However, this method still has a hard switching problem in the main PFC switch. The average switching loss in the main PFC switch due to turn-on and turn-off transitions can be approximated as follows:
where the switching loss with the FRD is Psw-FRD, the switching loss with the SiC is Psw-SiC, and fsw is the switching frequency of the PFC rectifier. This shows that the switching loss in the main PFC switch changes linearly according to fsw, IRR, and Iin.av. Fig. 7 shows the calculated conduction losses using (38) and (39). The typical values of tt, tf, QRR, and trr are obtained from the data-sheets provided by manufactures. From the results of Fig. 7, the switching losses of the PFC rectifier using the SiC diode are much smaller than those of the PFC rectifier using the FRD. However, the switching loss of the PFC rectifier using the SiC diode also increases according to the increment of the output power.
Fig. 7.Switching losses in the conventional FRD diode based PFC and SiC diode based PFC.
In order to increase the power conversion efficiency, a proposed lossless snubber circuit with the coupled inductor method is proposed in this paper. Fig. 6(b) shows the operating waveforms of the proposed lossless snubber PFC. When the main PFC switch turns on, the inductors of the proposed snubber limit the rising rate of the switch current. Therefore, current passing through the main PFC switch during the turn-on and turn-off intervals is reduced. The turn-on operation is similar to the mechanism of zero current switching and the turn-off operation is similar to the mechanism of zero voltage switching. Therefore, the switching loss during the turn-on and turn-off intervals will be dramatically reduced. The switching loss of the proposed PFC rectifier is negligible when compared to the cases of the conventional FRD diode and the SiC diode.
V. EXPERIMENTAL RESULTS
Fig. 8 shows a prototype converter of the proposed PFC boost rectifier. Table I shows the designed parameters of the proposed boost PFC converter. A 3.3kW (380V/8.7A) PFC boost rectifier using the proposed lossless snubber has been designed and evaluated at 220Vac input voltage. The fixed switching frequency was 50kHz and the PFC rectifier power stage is operated in the CCM using a commercial controller (L4981) made by STmicroelectonics. The boost inductor is designed to L1 = 800μH. The component values of the proposed snubber circuit are also given in Table I. In order to reduce the current stress caused by the reverse recovery characteristics, the coupled inductors are designed so that L2 = L3 = 3.6μH. The coupled inductors were implemented using a commercial toroidal core (CH610125, Chang-sung, two cores in parallel) with 32 turns for the boost inductor and 2 turns for each of the inductors. If the coupling coefficient in the inverse coupled windings case is defined to k = 1, the equivalent inductance is Leq = 14.4μH. Based on (27), the variation value of the reverse recovery current is determined by dIL2(t0)/dt = 26A/μS. It is lower than the recommended value of 100A/μS. The peak value of the limited reverse recovery current through the main switch, Irr, has been designed with a 30% design margin of the rated current.
Fig. 8.Prototype circuit of the proposed PFC.
By setting α = 50 in (29) and VCx(t3) = 200V in (30), the snubber capacitance Cx is found to be Cx ≥ 125nF. Therefore, the practical values of 2.2nF and 130nF are selected for Cs and Cx.
The measured waveforms of the input voltage and current are shown in Fig. 9 (a) and (b), respectively. Fig. 9 (a) shows the PFC operation of the prototype rectifier at 20% of the rated load condition. Fig. 9 (b) shows the input voltage and current waveforms under the rated load condition. In these figures, both the line voltage and current waveforms under a light load and a full load are almost in phase. From these results, the PFC operation of the prototype rectifier using the proposed lossless snubber is verified.
Fig. 9.The waveforms of the input voltage and current of the prototype PFC rectifier. (a) 660W output power, (b) 3.3kW rated output power.
Fig. 10 (a) and (b) show the switching waveforms of the boost PFC rectifier using a SiC diode. Fig. 10 (a) shows the switch voltage and current during the turn-on transition. The current and voltage spikes are smaller than when using the conventional FRD diode since the SiC diode has a very small reverse recovery charge [23]. However, it still has a hard switching problem of the main switch. It also has switching losses during the turn-off transition as shown in Fig. 10 (b). If he rated power or the switching frequency increases, the power loss of the SiC diode is also increased.
Fig. 10.Measured waveforms of the switch voltage and current using SiC diode under half load condition. (a) During turn-on transition, (b) During turn-off transition.
If the FRD is used as a boost diode without snubber circuits, it makes a high current spike and a high voltage spike because of the high reverse recovery current from the FRD diode when it turns off. Fig. 11 (a) shows the voltage and current waveforms of the main switch during the turn-on transition. The experimental results verify that the variation of the output diode current is limited by the proposed turn-on lossless snubber using coupled inductors. When the main switch S turns off, the reverse recovery current of the output diode is limited by the coupled inductors L2 and L3. Fig. 11 (b) shows the voltage and current waveforms of the main switch during the turn-off transition. The proposed turn-off lossless snubber (Cs, Ds1, and Ds2) can reduce the voltage stresses across the main switch. The proposed lossless snubber does not introduce extra voltage and current stresses on the main switch during the turn-on and turn-off periods. It reduces the overlapped area between the voltage and current of the switch. The soft switching using the proposed lossless snubber can increase the power conversion efficiency of the PFC rectifier and reduce the heat emission from the switching MOSFETs.
Fig. 11.Measured waveforms of the switch voltage and current using the proposed lossless snubber circuit under the rated load condition. (a) During turn-on transition. (b) During turn-off transition.
Fig. 12 shows the measured curve of the power factor according to the output power. Above the 40% power region, the PFC rectifier can achieve an almost unity power factor. From the experimental results, the PFC operation of the proposed prototype rectifier shows good performance under output load variations.
Fig. 12.Measured power factor of the prototype rectifier.
Fig. 13 shows the efficiency curve of the CCM PFC boost rectifier using the proposed lossless snubber. It is measured at an input voltage of 220Vac. In order to compare the power conversion efficiency, a commercial SiC diode (CS320060D) was used for the rectifier diode of the boost PFC rectifier. From the experimental results, it can be seen that the efficiency of the rectifier with the SiC diode is slightly higher than the efficiency of the proposed lossless snubber method under 40% of the rated load. The experimental results in Fig. 13 are slightly different from the calculation of the conduction loss in Fig. 7 because of neglecting the switching loss of the proposed PFC rectifier. However, at over 40% of the rated load, the power conversion efficiency of the FRD-based rectifier with the proposed lossless snubber circuit is higher than that of the SiC diode based rectifier. The increasing power losses of the SiC diode based rectifier are a hard switching problem of the main switch and the conduction losses of the SiC diode. The conduction loss of the rectifier diode depends on the forward voltage drop of the SiC diodes. The SiC diode (C3D20060D) has a higher forward voltage drop (Vf = 1.5) than that of the FRD DSEI2X101-06 (Vf = 1.17). At the rated load of 3.3 kW, the difference in the conduction losses between the lossless snubber circuit and the SiC diode in the PFC rectifier is 5W. Fig. 14 shows calculated power loss graphs at the rated load condition using (35), (38), and (39). As shown in Fig. 14, the switching losses of the conventional PFC rectifier using the FRD without snubber circuits is significantly increased. In addition, the proposed lossless snubber circuit can reduce more power loses than the SiC diode in this condition. The power conversion efficiency of the conventional PFC method is decreased by about 1.2% at 60% of the rated load when compared to the proposed methods. The power conversion efficiency of the proposed rectifier is higher than 97% when the output power is above 40% of the rated load. The highest efficiency is 97.48% at 2.64kW. At the rated output power of 3.3kW, the power conversion efficiency is 97.2%.
Fig. 13.Measured efficiency of the prototype rectifier.
Fig. 14.Power loss graph under the rated load condition.
VI. CONCLUSIONS
In this paper, the detailed performance improvement and the analysis of the operating effects of the coupled inductors were presented. In order to verify the improvement of the proposed snubber circuit, a 3.3kW prototype PFC rectifier using a commercial PFC controller was designed. The power conversion efficiency of the proposed circuit increases around 1.2% at 60% compared with the conventional methods. The power conversion efficiency of the proposed rectifier is higher than 97% when the output power is above 40% of the rated load. Even though the proposed technique requires a relatively large number of passive or active components, it is a good solution for high power rectifier applications by reducing the reverse recovery problems.
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