A Parallel Hybrid Soft Switching Converter with Low Circulating Current Losses and a Low Current Ripple

Abstract

A new parallel hybrid soft switching converter with low circulating current losses during the freewheeling state and a low output current ripple is presented in this paper. Two circuit modules are connected in parallel using the interleaved pulse-width modulation scheme to provide more power to the output load and to reduce the output current ripple. Each circuit module includes a three-level converter and a half-bridge converter sharing the same lagging-leg switches. A resonant capacitor is adopted on the primary side of the three-level converter to reduce the circulating current to zero in the freewheeling state. Thus, the high circulating current loss in conventional three-level converters is alleviated. A half-bridge converter is adopted to extend the ZVS range. Therefore, the lagging-leg switches can be turned on under zero voltage switching from light load to full load conditions. The secondary windings of the two converters are connected in series so that the rectified voltage is positive instead of zero during the freewheeling interval. Hence, the output inductance of the three-level converter can be reduced. The circuit configuration, operation principles and circuit characteristics are presented in detail. Experiments based on a 1920W prototype are provided to verify the effectiveness of the proposed converter.

I. INTRODUCTION

In medium and high power applications, three-phase AC/DC converters are usually used to provide stable DC voltages for industrial power units. Three-phase bridge or bridgeless power factor correctors (PFC) are normally adopted in the front stage to compensate the input power quality with a nearly unit power factor and low total harmonic distortion of the line current. Three-level DC/DC converters [1]-[8] are used as the second stage to provide a stable low output voltage with a high load current. Phase-shift pulse-width modulation (PSPWM) is normally used to generate gating signals and to regulate the output voltage. However, the main disadvantages of the conventional PSPWM full-bridge converter and three-level converters are a narrow ZVS range at the lagging-leg switches and high circulating current losses. Additional passives or active components added on the primary or secondary side have been presented in [9]-[14] to reduce the circulating current in the freewheeling state and to extend the ZVS range of the lagging-leg switches. The basic idea behind these techniques is to generate a positive voltage at the freewheeling state in order to reduce the primary current to zero. However, additional power losses can occur on these extra components. Thus, the total circuit efficiency cannot be improved. A large leakage inductance or low magnetizing inductance can be used on the primary side to extend the ZVS range of the lagging-leg switches. However, the large leakage inductor will also increase the duty cycle loss and decrease the effective duty cycle. Thus, the conduction losses on the primary side will be increased and the circuit efficiency is reduced.

A parallel soft switching DC/DC converter is presented in this paper. The proposed converter includes two hybrid circuit modules to reduce the current stress of the active and passive power components. Each circuit module includes a three-level PWM converter and a half-bridge converter to achieve the functions of a low circulating current, a wide ZVS range and a low output inductance. The half-bridge converter shares the lagging-leg switches of the three-level converter and operates at a 0.5 duty cycle control. The primary side current of the half-bridge converter can enable the lagging-leg switches to be turned on at ZVS from a light load. Thus, the narrow ZVS range in the conventional three-level converter is overcome in the proposed converter. The secondary windings of two converters are connected in series so that a positive rectified voltage instead of zero voltage is generated on the secondary side in the freewheeling state. This positive rectified voltage is reflected to the primary side to reduce the circulating current to zero. Thus, the circulating current losses on the primary side can be reduced and the circuit efficiency is improved. Since the output inductor voltage during the freewheeling state is also reduced, the size of output filter inductor can also be reduced. Two hybrid circuit modules are operated using the interleaved PWM scheme. The input and output current ripples can be reduced. During the active mode, both the three-level converter and half-bridge converter transfer energy from the input voltage to the output load. In the freewheeling state, only the half-bridge converter transfers energy to the output load. The circuit configuration, operation principles and circuit characteristics are presented in detail. Finally, experimental results with a 1920W laboratory prototype converting 800V to 48V/40A are provided to validate the theoretical analysis and effectiveness of the proposed converter.

 

II. PROPOSED CONVERTER

Fig. 1(a) shows a circuit diagram of the proposed hybrid DC/DC converter. The main proposed converter includes a three-level converter and a half-bridge converter. The lagging-leg switches S2 and S3 are shared by the three-level converter (Cin1, Cin2, Da, Db, Cf1, Cf2, S1-S4, T1, Lr1, Cr, D1, D4 and Lo) and the half-bridge converter (Cf1, Cf2, S2, S3, T2, Lr2, D2, D3 and Lo). The energy stored in the output inductor is reflected to the primary side to achieve the ZVS operation of all of the switches instead of only the leading-leg switches in the conventional three-level converter. The resonant capacitor Cr in the three-level converter is used to reduce the primary current to zero in the freewheeling state. Therefore, the circulating current losses on the primary side can be reduced. The secondary windings of the two converters are connected in series so that the rectified voltage vr is positive instead of zero in the freewheeling state. Hence, the inductor voltage vLo in the freewheeling state is reduced and the inductor current ripple is decreased when compared to the conventional three-level converter.

Fig. 1.The proposed hybrid converter (a) circuit diagram (b) main PWM waveforms.

 

III. OPERATION PRINCIPLES

In the proposed hybrid converter, the phase-shift PWM scheme is adopted to control the necessary duty cycle on the primary side and to regulate the output voltage. The PWM signal of S2 (S3) is phase-shifted with respective to the PWM signal of S1 (S4). S1 (S2) and S4 (S3) operate complementarily with a short dead time to avoid short circuits at the high voltage side. The main PWM waveforms of the proposed converter are shown in Fig. 1(b). It is assumed that the power semiconductors including S1-S4, Da-Dd and D1-D4 are ideal, the turns ratios of T1 and T2 are n1 and n2, respectively, CS1=CS2=CS3=CS4=Coss, and VCin1=VCin2=VCf1+VCf2=Vin/2. Based on the on/off states of S1-S4, Da-Dd and D1-D4, there are six operating stages in each half of a switching period. The duty cycle δ is defined as the turn-on interval of (S1 and S2) or (S3 and S4). Fig. 2 gives the topological circuits for the six operating stages during the first half of a switching period.

Fig. 2.Operation modes 1-6 of the proposed converter during the first half switching period (a) mode 1 (b) mode 2 (c) mode 3 (d) mode 4 (e) mode 5 (f) mode 6.

Stage 1 [t0 - t1]: Prior to time t0, switches S1 and S2 and diodes D1 and D3 are conducting. The primary currents iLr1 and iLr2 are positive. At time t0, the diode current iD3 decreases to zero. The voltages vab=Vin/2 and vac=VCf1=Vin/4. The primary currents can be approximately expressed as iLr1 ≈ Io/n1 and iLr2 ≈ Io/n2. The capacitor voltage vCr is given as vCr (t) ≈ vCr (t0 ) + Io (t - t0 ) /(Crn1) . The voltage variation ΔvCr in this stage can be expressed as:

Therefore, the rectified voltage at the transformer secondary side is given as:

The output inductor voltage vLo=vr-Vo>0 so that the inductor current iLo increases in this stage.

Stage 2 [t1 - t2]: At time t1, switch S1 is turned off. The energy stored in the output inductor Lo is reflected to the primary side to charge CS1 and discharge CS4.

Thus, S4 can be easily turned on under ZVS and the ZVS condition of S4 is given as:

At t2, capacitor CS4 is discharged to zero voltage and the rectified voltage vr is decreased to Vin/(4n2). The time interval Δt12 in stage 2 can be obtained from (4) and expressed as:

The dead time td between switches S1 and S4 must be greater than the time interval Δt12 in order to turn on S4 under ZVS.

Stage 3 [t2 - t3]: At time t2, vCS4=0. Since iS4(t2)<0, the anti-parallel diode of S4 conducts. S4 can be turned on at this moment under ZVS. The voltages vab=0 and vac=Vin/4 and diodes D1 and D2 are conducting to commutate the secondary side current of T1. The diode current iD1 decreases and iD2 increases. In this stage, Lr1 and Cr are resonant to reduce the primary current iLr1 to zero.

Stage 4 [t3 - t4]: At time t3, the diode current iD1 is decreased to zero and only diode D2 conducts the load current. The primary current iLr1 of the three-level converter is decreases to zero so that no power is transferred through the three-level converter. The circulating current in this freewheeling state is reduced to zero and there is no circulating loss in this stage. The primary current of the half-bridge converter iLr2=iLo/n2 and the energy is transferred from the input capacitor Cf1 to the output load by the half-bridge converter in this stage. The rectified voltage vr ≈ Vin/(4n2). The output inductor voltage vLo ≈ Vin/(4n2)-Vo<0. Therefore, the inductor current iLo decreases in this stage.

Stage 5 [t4 - t5]: Switch S2 is turned off at t4. Since iLr2(t4)>0 and D2 is conducting, the energy stored in Lo is reflected to the primary sides to charge CS2 and discharge CS3.

The ZVS condition of S3 is approximately given as:

At time t5, CS3 is discharged to zero voltage. The time interval Δt45is obtained as:

The dead time td between switches S2 and S3 must be greater than the time interval Δt45 in order to turn on S3 under ZVS.

Stage 6 [t5 - t6]: At time t5, CS3 is discharged to zero voltage. Since iS3(t5)<0, the anti-parallel diode of S3 conducts. Thus, S3 can be turned on at this moment under ZVS. The voltages vab=-Vin/2 and vac=-Vin/4. D2 and D4 are both conducting to commutate the output inductor current iLo. Since D2 and D4 are conducting, it is possible to derive that the secondary winding voltages of T1 and T2 are vT2,s=-0.5vT1,s. The secondary winding voltage vT1,p (t5 ) is approximately equal to -Vin / 2 - IoδTs /(2Crn1). Thus, the primary winding voltage of vT2,p at time t5 can be given as:

The primary current iLr2 in this stage is decreased and is expressed as:

The primary current iLr1 is decreased from zero to -n1Io, and the primary current iLr2 is also decreased from n2Io to -n2Io. At time t6, the diode current iD2 is decreased to zero and only diode D4 conducts the load current. Then, the circuit operations in the first half switching period are complete.

 

IV. PARALLEL HYBRID CONVERTER

In order to provide more power to the output load, increase the ripple frequency and reduce the input and output current ripple, a parallel hybrid converter with the interleaved PWM scheme is proposed and its circuit diagram is shown in Fig. 3(a). There are two circuit modules in the proposed parallel hybrid converter. Each circuit module supplies one-half of the rated power to the output load. The interleaved PWM scheme is adopted to generate the necessary gating signals for the power switches. The PWM signals of S5~S8 are phase-shifted by one-fourth of a switching period with respective to the PWM signals of S1~S4. In each of the circuit modules, the phase-shift PWM scheme is used to control the necessary duty cycle on the primary side and to regulate the output voltage. Fig. 3(b) shows the key PWM waveforms of the proposed parallel hybrid converter during one switching period. The PWM waveforms of the two circuit modules are similar and they are phase-shifted with one-fourth of a switching period. The circuit operations of each circuit module were discussed in the previous section. The main advantages of the interleaved PWM operation of the proposed parallel converter include less current ripple at the output side, double the switching frequency, and less the current stress of the passive components at the high current side. The circuit components of the two modules are identical and the duty cycle of each circuit module is the same due to the controller. Thus, current sharing of each circuit module can be achieved. Even if the circuit parameters of each module have some variation, the output inductor currents are almost balanced.

Fig. 3.Proposed parallel hybrid converter (a) circuit diagram (b) main PWM waveforms.

 

V. CONVERTER PERFORMANCE ANALYSIS

Since the duration periods of stages 2, 3 and 5 are very narrow when compared to the periods of the other stages, only stages 1, 4 and 6 are considered to derive the DC voltage gain. In stage 6, the rectified voltage vr1 = vT2,p (t5 ) . The duty loss in stage 6 is related to the load current and the resonant inductance Lr2. This duty loss δ6 exists in the conventional three-level converter and the proposed converter. The average rectified voltage Vr in stage 1 is approximately equal to Vin/(2n1)+Vin/(4n2) with the time interval δeffTs, where δeff= δ-δ6. In stage 4, the average rectified voltage Vr is equal to Vin/(4n2) with the time interval (0.5-δeff)Ts. The flux balance on the output inductor Lo can derive the output voltage Vo.

However, the output voltage of the conventional three-level converter is expressed as:

where n is the turns ratio of the conventional three-level PWM converter. Based on (13) and (14), a comparison of the normalized voltage conversion ratios of the proposed converter and the conventional three-level converter is given as:

From (15), the proposed converter has a higher voltage gain compared to the conventional three-level converter. The turns ratio n1 of the proposed converter can be larger than that in the conventional three-level converter with the same effective duty ratio. The larger turns ratio n1 will decrease the conduction losses and also reduce the voltage stress of the rectifier diodes. The power ratings and power percentage of the three-level converter and half-bridge converter in the proposed circuit are given as:

It can be seen that most of the load power is delivered by the half-bridge converter at a low effective duty cycle. In Fig. 3, the voltage stress of the rectifier diodes can be approximately obtained if the voltage spike on the diodes is neglected.

In the conventional three-level converter, the voltage stress of the rectifier diodes is vstress,D,convention al > Vo / δeff . Therefore, the voltage stress of the rectifier diodes in the proposed converter in (20) and (21) is less than the voltage stress in the conventional three-level converter with the same duty ratio. During the freewheeling interval in stage 4, the output inductor voltage vLo is equal to Vin/(4n2)-Vo with the duration period (0.5-δeff)Ts. Thus, the output inductor value can be derived as:

Based on (13) and (22), the necessary output inductance Lo is expressed as:

In the conventional three-level converter, the output inductance is given as:

The output inductance ratio between the proposed converter and the conventional three-level converter is given as:

This means that the proposed converter has less output inductance based on the same current ripple and effective duty cycle.

 

VI. EXPERIMENTAL RESULTS

The proposed parallel hybrid soft switching DC/DC converter is designed and implemented in this section to confirm the theoretical analysis and to verify the effectiveness of the proposed converter. In the proposed converter, the input voltage Vin is from 750V to 800V. The output voltage Vo is 48V and the rated load current Io,max is 40A. The switching frequency of the active switches is 100kHz. The two circuit modules are operated with the interleaved phase-shift PWM scheme. Each circuit module provides one-half of the load current to the output load, i.e. ILo1=ILo2=Io,max/2=20A. IRFP460 MOSFETs were used for switches S1~S8. VF30200C and MUR860 fast recovery diodes were used for D1~D8 and Da~Dd, respectively. The input split capacitances Cin1 and Cin2 are 360 μF. The flying capacitances Cf1~Cf4 are 1 μF. The resonant capacitances Cr1 and Cr2 are 18nF. The output capacitance Co is 4000 μF. The turns ratios of T1 and T3 are 54 turns: 4 turns: 4 turns. The turns ratios of T2 and T4 are 27 turns: 4 turns: 4 turns. The leakage inductances Lr1 and Lr3 are 14 μH, and the leakage inductances Lr2 and Lr4 are 22 μH. The output inductances Lo1 and Lo2 are 5 μH. The ZVS range for all of the switches in the proposed converter is from 25% load to full load. Fig. 4 shows a photograph of the proposed converter. The gate voltages of S1~S8 for different load conditions and an 800V input voltage are shown in Fig. 5. The PWM signals of S5~S8 are phase-shifted one-fourth of a switching period with respect to the PWM signals of S1~S4. Fig. 6 shows the measured gate voltage and drain voltage of switches S1~S8 at a 25% load. It is clear that all of the switches are turned on under ZVS from a 25% load. Figs. 7 and 8 show the primary side voltages and currents of the proposed converter at a 50% load and a 100% load, respectively. There are three voltage levels on the voltages vao and vco and two voltage levels on the voltages vab and vcd. The circulating currents iLr1 and iLr3 of the three-level converters are all reduced to zero in the freewheeling state (vab=vcd=0). Therefore, the conduction losses on the primary side in the freewheeling state are reduced. The primary currents iLr3 and iLr4 of circuit module 2 are phase shifted with respectively to the primary currents iLr1 and iLr2 of circuit module 1, respectively. Fig. 9 shows the measured results of the secondary side voltage and currents of circuit module 1 at full load. It can be seen that diodes D2 and D3 are conducting during the freewheeling interval. Fig. 10 shows the measured waveforms of the output inductor currents at a 100% load. The output inductor current iLo2 is phase shifted to the output inductor current iLo1. It is clear that the resulting output current ripple Δ(iLo1+iLo2) is less than the inductor ripple currents ΔiLo1 and ΔiLo2 due to the interleaved PWM scheme. The measured circuit efficiencies of the proposed converter are 93.5%, 95.7% and 93.3% at a 25% load, a 50% load and a 100% load, respectively. However, the measured circuit efficiencies of the conventional three-level converter under the same power ratings are 91.9%, 93.4% and 92.1% at a 25% load, a 50% load and a 100% load, respectively. Based on the measured results, it is clear that the proposed converter has better circuit efficiency compared to the conventional three-level converter.

Fig. 4.Photograph of the prototype circuit (a) analog control board (b) main power circuit.

Fig. 5.Measured PWM waveforms of S1~S8) at (a) Po=960W (50% load) (b) Po=1920W (100% load).

Fig. 6.Measured gate and drain voltages of S1~S8 at 25% load (a) S1 and S5 (b) S2 and S6 (c) S3 and S7 (d) S4 and S8.

Fig. 7.Measured primary side voltage and current waveforms at 50% load (Po=960W).

Fig. 8.Measured primary side voltage and current waveforms at full load (Po=1920W).

Fig. 9.Measured secondary side voltage and current waveforms of circuit module 1 at full load (Po=1920W).

Fig. 10.Measured output inductor currents at full load (Po=1920W).

 

VII. CONCLUSION

A new parallel hybrid soft switching converter is presented for medium voltage and power applications. Two circuit modules operated with the interleaved PWM scheme are used in the proposed converter to lessen the current stresses of the active and passive components and to reduce the input and output current ripples. In each circuit module, one three-level converter and one half-bridge converter sharing the lagging-leg switch are used to achieve a wide ZVS range for all of the switches, a low circulating current on the primary side, and less output inductor current ripple. Therefore, the circuit efficiency in the proposed converter is higher than the circuit efficiency in the conventional three-level converter. The circuit configuration, operation principles and circuit characteristics are discussed in detail. Experimental results based on a 1920W prototype confirm the effectiveness of the proposed converter.